The present invention relates to digital communication where Orthogonal Frequency Division Multiplexing (OFDM) is employed. More particularly, the invention relates to channel estimation in an OFDM communications system.
OFDM is a method that has been increasingly popular for transmitting digital information. Currently it is, for example, used for Digital Audio Broadcasting (DAB), Digital Video Broadcasting (DVB) such as DVB-Terrestrial (DVB-T), and for some Wireless Local Area Network (WLAN) standards like IEEE 802.11a and IEEE 802.11g. One of the reasons for using OFDM is that it allows for communication over highly time-dispersive channels using reasonable complexity at the receiver side.
In an OFDM system, a number of sub-carriers (henceforth referred to simply as “carriers”) are independently modulated, each by its own data. The modulation can be in accordance with a number of well-known techniques, such as Quadrature Amplitude Modulation (QAM) or n-ary Phase Shift Keying (n-PSK). The baseband signal in an OFDM system is then the sum of these modulated sub-carriers. The baseband signal is then used to modulate a main radio frequency (RF) signal. An important aspect of demodulating such a signal (thereby retrieving the underlying baseband signal) involves processing it by a Fast Fourier Transform (FFT).
Whether a channel should be considered highly time-dispersive or not depends on the symbol rate that is used by the system. As a rule-of-thumb, a channel might be considered as non-dispersive if the root mean square (rms) delay spread of the channel is less than 10% of the symbol duration. Thus advantages of OFDM become more pronounced as the supported data rate is increased, which is exactly the case for most of the emerging systems.
The way to handle large delay spreads for a system based on OFDM is to make use of a guard interval (GI). The GI (also referred to in the literature as a “cyclic prefix”, or “CP”) is simply a copy of the last part of an OFDM symbol that is sent before the actual symbol. This is schematically illustrated in FIG. 1, which shows a number of symbols. An exemplary one of the symbols 101 includes a last portion 103 that is transmitted as a preceding guard interval 105 (time flows from left to right in the figure). Other guard intervals are similarly formed from end portions of their immediately succeeding symbols.
It is well-known that for a system based on OFDM the effect of the time-dispersive channel, known as inter-symbol interference (ISI), can be avoided provided that the length of the GI, TG, is at least as long as the duration of the impulse response of the channel. The term Tm is henceforth used herein to denote the maximum delay spread, as opposed to, for example, a root-mean-square (rms) delay spread value. Because of the ability of an OFDM system to handle large delay spreads, it is very suitable for so-called Single Frequency Networks (SFN), which might be used for broadcasting. (In a single frequency network, geographically spaced transmitters operate on a same frequency. To reduce interference, they are time synchronized with one another.)
Suppose that the information carrying part of the OFDM symbol begins at t=0, and that the length of the guard interval is TG. If the channel has a maximum delay spread, Tm, the requirement on the start of the FFT window is given by−TG+Tm≦t≦0.  (1)
Thus, as long as Tm≦TG it is possible to avoid ISI if t is chosen according to equation (1). However, if Tm>TG the issue is to choose t such that the effect of ISI is minimized. For systems designed for use in a SFN, the guard interval is typically so large that the first situation is the likelier one.
Now, as discussed above, ISI free reception is possible whenever Tm≦TG. However, this requires identifying the start of the information carrying part of the signal. For this reason, OFDM receivers include arrangements for estimating the timing and frequency of the received signal.
To further improve performance, OFDM receivers typically include channel estimators, whose job is to dynamically determine the channel response. This information is then used to enable the receiver to process the received signal in a way that compensates for the time dispersion effects of the channel.
A conventional way of determining the channel response in an OFDM receiver is to dedicate certain ones of the carriers for use in conveying pilot signals. The pilot signals contain known information that permits the channel estimator to determine the channel response on that carrier frequency at that particular instant in time by comparing the actually received signal with an expected signal (i.e., one that the receiver would have expected to receive under ideal channel conditions). The carriers conveying the pilot signals are spaced apart in frequency by an amount that permits the channel response of carriers lying in-between the pilot carriers to be accurately estimated by interpolating the channel responses determined for the pilot carriers.
FIG. 2 is a block diagram of an exemplary OFDM receiver. An analog signal, r(t), generated by receiving and downconverting to baseband a radiofrequency signal, is supplied to an analog-to-digital (A/D) converter 201. The downconversion from the radio frequency might alternatively be performed in several steps, so that the signal at the input of the A/D converter 201 is at an intermediate frequency (IF), and where the down-conversion from IF to baseband is done just after the A/D converter, prior to the further processing now to be described.
The digitized signal, r(k), is then supplied to a coarse timing and frequency estimation unit 203, which generates a coarse estimate of the timing and frequency offset of the received signal. (The frequency offset is the difference between the frequency of the transmitted signal and the frequency of the received signal.) This information is supplied to a frequency correction unit 205 as well as a GI removal unit 207. The GI removal unit 207 also receives the output of the frequency correction unit 205. Based on the best timing and frequency information available, the GI removal unit 207 removes the GI and supplies the information part of the received signal to an FFT unit 209, whose output is supplied to the remainder of the receiver, including a refined timing and frequency estimation unit 211, which is able to generate more accurate timing and frequency information from the FFT output signal. The more accurate frequency information is fed back to the frequency correction unit 205 to improve the receiver's performance. The more accurate timing information is similarly fed back to the GI removal unit 207 to improve the receiver's performance.
The output of the FFT unit 209 is also supplied to a channel estimator 213, which generates a complete estimate of the channel response, as explained above.
Although any value of t that fulfills Eq. (1) will ensure that inter-symbol interference (ISI) is avoided, the choice of t might still have an impact on a receiver's performance. Specifically, it can be shown that if the start of the FFT window is placed at ε samples too early (ε≧0), then the effect on the kth carrier at the output of the FFT,  will be=X(k)e−i2πkε/N,  (2)where X(k) is what the FFT output would be if ε=0, N is the number of samples corresponding to the duration of the information carrying part of one symbol, and k is an index representing the position of a carrier.
Thus, the different carriers will be rotated differently depending on their position, as represented by the index k. As a result, when interpolation is done in the frequency domain (i.e., when the channel response for carriers containing data is estimated using carriers containing pilots), performing the interpolation will become unnecessarily difficult. Consequently, although the same performance might be obtained even if the FFT window is placed ε samples too early, it usually comes at the expense of requiring a more complicated channel estimation procedure.
Since the effect of placing the FFT window ε samples too early is known, it can be easily compensated for simply by multiplying by ei2πkε/N prior to performing channel estimation. This is well-known, and described for instance in A. Palin and J. Rinne, “Enhanced symbol synchronization method for OFDM system in SFN channels,” Globecom'98, Sydney, pp. 2788-2793. Thus, it is well-known that one might compensate for a non-optimal placement of the FFT window by properly rotating the signal before doing the channel estimation. However, given that ε is known, one might as well move the FFT window in the optimal position, and by that avoid the extra computations that would be required if the compensation were done after the FFT.
With an optimally chosen position for the FFT window, the required complexity for interpolating in frequency depends on the delay spread of the channel. Thus, if the delay spread of the channel is estimated, this knowledge can be used not only to design a (Wiener) filter, but it can also be used to determine how complicated the filter has to be. This is described in detail in U.S. patent application Ser. No. 10/920,928, entitled “Channel Estimation by Adaptive Interpolation” by Leif Wilhelmsson et al. A conclusion made in U.S. Ser. No. 10/920,928 is that to design an optimum filter, it is not sufficient to estimate only the delay spread (for instance the rms delay spread or the maximum delay spread); the correlation function (in the frequency direction) for the channel is also required.
Since the channel estimation is one of the most critical parts in an OFDM system for obtaining good performance, it is very important for the interpolation performance to be of sufficiently high quality. In addition, since the channel estimation accounts for a significant part of the computational complexity, it is also important to reduce the complexity of the interpolation filters that are used.